Method and system for weight determination in a single channel (SC) multiple-input multiple-output (MIMO) system for WCDMA/HSDPA

ABSTRACT

In wireless systems, method and system for weight determination in a single channel (SC) multiple-input multiple-output (MIMO) system for WCDMA/HSDPA are provided. Models of signals received in multiple receive antennas may be determined in a single weight baseband generator (SWBBG) from propagation channel estimates and noise statistics. The models may be utilized to determine combined signal and noise components. The combined signal and noise components may be utilized to determine a plurality of signal-to-noise ratio (SNR) or signal-to-interference-and-noise ratio (SINR) values for various phase and/or amplitude factors. The SINR may be utilized when either single or multiple interfering signals are present. A highest of the SINR or SNR values may be selected to determine a channel weight to apply to the additional receive antennas.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This patent application makes reference to, claims priority to andclaims benefit from U.S. Provisional Patent Application Ser. No.60/616,298 filed on Oct. 6, 2004.

This application makes reference to:

-   U.S. patent application Ser. No. 11/173,870 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/174,303 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,502 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,871 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,964 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,252 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,756 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,305 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,759 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,689 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,304 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,129 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,779 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,702 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,727 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,726 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,781 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/174,067 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,854 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,911 filed Jun. 30, 2005; and-   U.S. patent application Ser. No. 11/174,403 filed Jun. 30, 2005.

The above referenced applications are hereby incorporated herein byreference in their entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to the processing ofwireless communication signals. More specifically, certain embodimentsof the invention relate to a method and system for weight determinationin a single channel (SC) multiple-input multiple-output (MIMO) systemfor WCDMA/HSDPA.

BACKGROUND OF THE INVENTION

Mobile communications has changed the way people communicate and mobilephones have been transformed from a luxury item to an essential part ofevery day life. The use of mobile phones is today dictated by socialsituations, rather than hampered by location or technology. While voiceconnections fulfill the basic need to communicate, and mobile voiceconnections continue to filter even further into the fabric of every daylife, the mobile Internet is the next step in the mobile communicationrevolution. The mobile Internet is poised to become a common source ofeveryday information, and easy, versatile mobile access to this datawill be taken for granted.

Third generation (3G) cellular networks have been specifically designedto fulfill these future demands of the mobile Internet. As theseservices grow in popularity and usage, factors such as cost efficientoptimization of network capacity and quality of service (QoS) willbecome even more essential to cellular operators than it is today. Thesefactors may be achieved with careful network planning and operation,improvements in transmission methods, and advances in receivertechniques. To this end, carriers need technologies that will allow themto increase downlink throughput and, in turn, offer advanced QoScapabilities and speeds that rival those delivered by cable modem and/orDSL service providers. In this regard, networks based on wideband CDMA(WCDMA) technology may make the delivery of data to end users a morefeasible option for today's wireless carriers.

FIG. 1A is a technology timeline indicating evolution of existing WCDMAspecification to provide increased downlink throughput. Referring toFIG. 1A, there is shown data rate spaces occupied by various wirelesstechnologies, including General Packet Radio Service (GPRS) 100,Enhanced Data rates for GSM (Global System for Mobile communications)Evolution (EDGE) 102, Universal Mobile Telecommunications System (UMTS)104, and High Speed Downlink Packet Access (HSDPA) 106.

The GPRS and EDGE technologies may be utilized for enhancing the datathroughput of present second generation (2G) systems such as GSM. TheGSM technology may support data rates of up to 14.4 kilobits per second(Kbps), while the GPRS technology, introduced in 2001, may support datarates of up to 115 Kbps by allowing up to 8 data time slots per timedivision multiple access (TDMA) frame. The GSM technology, by contrast,may allow one data time slot per TDMA frame. The EDGE technology,introduced in 2003, may support data rates of up to 384 Kbps. The EDGEtechnology may utilizes 8 phase shift keying (8-PSK) modulation forproviding higher data rates than those that may be achieved by GPRStechnology. The GPRS and EDGE technologies may be referred to as “2.5G”technologies.

The UMTS technology, introduced in 2003, with theoretical data rates ashigh as 2 Mbps, is an adaptation of the WCDMA 3G system by GSM. Onereason for the high data rates that may be achieved by UMTS technologystems from the 5 MHz WCDMA channel bandwidths versus the 200 KHz GSMchannel bandwidths. The HSDPA technology is an Internet protocol (IP)based service, oriented for data communications, which adapts WCDMA tosupport data transfer rates on the order of 10 megabits per second(Mbits/s). Developed by the 3G Partnership Project (3GPP) group, theHSDPA technology achieves higher data rates through a plurality ofmethods. For example, many transmission decisions may be made at thebase station level, which is much closer to the user equipment asopposed to being made at a mobile switching center or office. These mayinclude decisions about the scheduling of data to be transmitted, whendata is to be retransmitted, and assessments about the quality of thetransmission channel. The HSDPA technology may also utilize variablecoding rates. The HSDPA technology may also support 16-level quadratureamplitude modulation (16-QAM) over a high-speed downlink shared channel(HS-DSCH), which permits a plurality of users to share an air interfacechannel

In some instances, HSDPA may provide a two-fold improvement in networkcapacity as well as data speeds up to five times (over 10 Mbit/s) higherthan those in even the most advanced 3G networks. HSDPA may also shortenthe roundtrip time between network and terminal, while reducingvariances in downlink transmission delay. These performance advances maytranslate directly into improved network performance and highersubscriber satisfaction. Since HSDPA is an extension of the GSM family,it also builds directly on the economies of scale offered by the world'smost popular mobile technology. HSDPA may offer breakthrough advances inWCDMA network packet data capacity, enhanced spectral and radio accessnetworks (RAN) hardware efficiencies, and streamlined networkimplementations. Those improvements may directly translate into lowercost-per-bit, faster and more available services, and a network that ispositioned to compete more effectively in the data-centric markets ofthe future.

The capacity, quality and cost/performance advantages of HSDPA yieldmeasurable benefits for network operators, and, in turn, theirsubscribers. For operators, this backwards-compatible upgrade to currentWCDMA networks is a logical and cost-efficient next step in networkevolution. When deployed, HSDPA may co-exist on the same carrier as thecurrent WCDMA Release 99 services, allowing operators to introducegreater capacity and higher data speeds into existing WCDMA networks.Operators may leverage this solution to support a considerably highernumber of high data rate users on a single radio carrier. HSDPA makestrue mass-market mobile IP multimedia possible and will drive theconsumption of data-heavy services while at the same time reducing thecost-per-bit of service delivery, thus boosting both revenue andbottom-line network profits. For data-hungry mobile subscribers, theperformance advantages of HSDPA may translate into shorter serviceresponse times, less delay and faster perceived connections. Users mayalso download packet-data over HSDPA while conducting a simultaneousspeech call.

HSDPA may provide a number of significant performance improvements whencompared to previous or alternative technologies. For example, HSDPAextends the WCDMA bit rates up to 10 Mbps, achieving higher theoreticalpeak rates with higher-order modulation (16-QAM) and with adaptivecoding and modulation schemes. The maximum QPSK bit rate is 5.3 Mbit/sand 10.7 Mbit/s with 16-QAM. Theoretical bit rates of up to 14.4 Mbit/smay be achieved with no channel coding. The terminal capability classesrange from 900 kbit/s to 1.8 Mbit/s with QPSK modulation, and 3.6 Mbit/sand up with 16-QAM modulation. The highest capability class supports themaximum theoretical bit rate of 14.4 Mbit/s.

However, implementing advanced wireless technologies such as WCDMAand/or HSDPA may still require overcoming some architectural hurdles.For example, the RAKE receiver is the most commonly used receiver inCDMA systems, mainly due to its simplicity and reasonable performanceand WCDMA Release 99 networks are designed so that RAKE receivers may beused. A RAKE receiver contains a bank of spreading sequence correlators,each receiving an individual multipath. A RAKE receiver operates onmultiple discrete paths. The received multipath signals may be combinedin several ways, from which maximum ratio combining (MRC) is preferredin a coherent receiver. However, a RAKE receiver may be suboptimal inmany practical systems, for example, its performance may degrade frommultiple access interference (MAI), that is, interference induced byother users in the network.

In the case of a WCDMA downlink, MAI may result from inter-cell andintracell interference. The signals from neighboring base stationscompose intercell interference, which is characterized by scramblingcodes, channels and angles of arrivals different from the desired basestation signal. Spatial equalization may be utilized to suppressinter-cell interference. In a synchronous downlink application,employing orthogonal spreading codes, intra-cell interference may becaused by multipath propagation. Due to the non-zero cross-correlationbetween spreading sequences with arbitrary time shifts, there isinterference between propagation paths (or RAKE fingers) afterdespreading, causing MAI. The level of intra-cell interference dependsstrongly on the channel response. In nearly flat fading channels, thephysical channels remain almost completely orthogonal and intra-cellinterference does not have any significant impact on the receiverperformance. On the other hand, the performance of the RAKE receiver maybe severely deteriorated by intra-cell interference in frequencyselective channels. Frequency selectivity is common for the channels inWCDMA networks.

Due to the difficulties faced when non-linear channel equalizers areapplied to the WCDMA downlink, detection of the desired physical channelwith a non-linear equalizer may result in implementing an interferencecanceller or optimal multi-user receiver. Both types of receivers may beprohibitively complex for mobile terminals and may require informationnot readily available at the mobile terminal. Alternatively, the totalbase station signal may be considered as the desired signal. However,non-linear equalizers rely on prior knowledge of the constellation ofthe desired signal, and this information is not readily available at theWCDMA terminal. The constellation of the total base station signal, thatis, sum of all physical channels, is a high order quadrature amplitudemodulation (QAM) constellation with uneven spacing. The spacing of theconstellation changes constantly due to transmission power control (TPC)and possible power offsets between the control data fields,time-multiplexed to the dedicated physical channels. The constellationorder may also frequently change due to discontinuous transmission. Thismakes an accurate estimation of the constellation very difficult.

In this regard, the use of multiple transmit and/or receive antennas mayresult in an improved overall system performance. These multi-antennaconfigurations, also known as smart antenna techniques, may be utilizedto mitigate the negative effects of multipath and/or signal interferenceon signal reception. It is anticipated that smart antenna techniques maybe increasingly utilized both in connection with the deployment of basestation infrastructure and mobile subscriber units in cellular systemsto address the increasing capacity demands being placed on thosesystems. These demands arise, in part, from a shift underway fromcurrent voice-based services to next-generation wireless multimediaservices that provide voice, video, and data communication.

The utilization of multiple transmit and/or receive antennas is designedto introduce a diversity gain and to suppress interference generatedwithin the signal reception process. Such diversity gains improve systemperformance by increasing received signal-to-noise ratio, by providingmore robustness against signal interference, and/or by permittinggreater frequency reuse for higher capacity. In communication systemsthat incorporate multi-antenna receivers, a set of M receive antennasmay be utilized to null the effect of (M−1) interferers, for example.Accordingly, N signals may be simultaneously transmitted in the samebandwidth using N transmit antennas, with the transmitted signal thenbeing separated into N respective signals by way of a set of N antennasdeployed at the receiver. Systems that utilize multiple transmit andreceive antennas may be referred to as multiple-input multiple-output(MIMO) systems. One attractive aspect of multi-antenna systems, inparticular MIMO systems, is the significant increase in system capacitythat may be achieved by utilizing these transmission configurations. Fora fixed overall transmitted power, the capacity offered by a MIMOconfiguration may scale with the increased signal-to-noise ratio (SNR).For example, in the case of fading multipath channels, a MIMOconfiguration may increase system capacity by nearly M additionalbits/cycle for each 3-dB increase in SNR.

However, the widespread deployment of multi-antenna systems in wirelesscommunications, particularly in wireless handset devices, has beenlimited by the increased cost that results from increased size,complexity, and power consumption. Providing separate RF chain for eachtransmit and receive antenna is a direct factor that increases the costof multi-antenna systems. Each RF chain generally comprises a low noiseamplifier (LNA), a filter, a downconverter, and an analog-to-digitalconverter (A/D). In certain existing single-antenna wireless receivers,the single required RF chain may account for over 30% of the receiver'stotal cost. It is therefore apparent that as the number of transmit andreceive antennas increases, the system complexity, power consumption,and overall cost may increase. This poses problems for mobile systemdesigns and applications.

Further limitations and disadvantages of conventional and traditionalapproaches will become apparent to one of skill in the art, throughcomparison of such systems with some aspects of the present invention asset forth in the remainder of the present application with reference tothe drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or system for weight determination in a single channel (SC)multiple-input multiple-output (MIMO) system for WCDMA/HSDPA,substantially as shown in and/or described in connection with at leastone of the figures, as set forth more completely in the claims.

These and other advantages, aspects and novel features of the presentinvention, as well as details of an illustrated embodiment thereof, willbe more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1A is a technology timeline indicating the evolution of theexisting WCDMA specification to provide increased downlink throughput.

FIG. 1B illustrates an exemplary HSDPA distributed architecture thatachieves low delay link adaptation, in connection with an embodiment ofthe invention.

FIG. 1C illustrates an exemplary Layer 1 HARQ control situated in a basestation to remove retransmission-related scheduling and storing from theradio network controller, in connection with an embodiment of theinvention.

FIG. 1D is a chart illustrating exemplary average carried loads forHSDPA-based macrocell and microcell systems, in connection with anembodiment of the invention.

FIG. 2A is a block diagram of an exemplary one-transmit (1-Tx) andmultiple-receive (M-Rx) antennas wireless communication system withreceiver channel weight generation, in accordance with an embodiment ofthe invention.

FIG. 2B is a block diagram of an exemplary two-transmit (2-Tx) andmultiple-receive (M-Rx) antennas wireless communication system withreceiver channel weight generation, in accordance with an embodiment ofthe invention.

FIG. 3A is a flow diagram illustrating exemplary steps for channelestimation in a 2-Tx and M-Rx antennas wireless communication system, inaccordance with an embodiment of the invention.

FIG. 3B illustrates an exemplary periodic phase rotation for an in-phase(I) signal received in one of the additional receive antennas, inaccordance with an embodiment of the invention.

FIG. 4A is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and 2-Rx antennassystem, in accordance with an embodiment of the invention.

FIG. 4B is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and M-Rx antennassystem, in accordance with an embodiment of the invention.

FIG. 4C is a block diagram of an exemplary RF phase and amplitudecontroller, in accordance with an embodiment of the invention.

FIG. 5 is a flow diagram illustrating exemplary steps in the operationof the single weight baseband generator (SWBBG) that may be utilized forchannel weight generation in a 2-Tx and M-Rx antennas system, inaccordance with an embodiment of the invention.

FIG. 6 is a flow diagram illustrating exemplary steps for determiningchannel weights in additional receive antennas utilizing signal-to-noiseratio (SNR) or signal-to-noise-to-interference ratio (SINR), inaccordance with an embodiment of the invention.

FIG. 7 is a flow diagram illustrating exemplary steps for determiningchannel weights by monitoring baseband combined channel estimates duringphase rotation, in accordance with an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method and systemfor weight determination in a single channel (SC) multiple-inputmultiple-output (MIMO) system for WCDMA/HSDPA. Propagation channelsreceived in multiple receive antennas may be estimated in a singleweight baseband generator (SWBBG) from combined propagation channelestimates. The channel estimates, interfering channel estimates, andestimated noise components may be utilized to determine a plurality ofsignal-to-noise ratio (SNR) or signal-to-interference-and-noise ratio(SINR) values for various phase and/or amplitude factors. The SINR maybe utilized when either single or multiple interfering signals arepresent. A highest of the SINR or SNR values may be selected todetermine a channel weight to apply to the additional receive antennas.This approach may provide a good compromise between implementationcomplexity and performance gains in the design and operation of MIMOsystems.

FIG. 1B illustrates an exemplary HSDPA distributed architecture thatachieves low delay link adaptation, in connection with an embodiment ofthe invention. Referring to FIG. 1B, there is shown terminals 110 and112 and a base station (BS) 114. HSDPA is built on a distributedarchitecture that achieves low delay link adaptation by placing keyprocessing at the BS 114 and thus closer to the air interface asillustrated. Accordingly, the MAC layer at the BS 114 is moved fromLayer 2 to Layer 1, which implies that the systems may respond in a muchfaster manner with data access. Fast link adaptation methods, which aregenerally well established within existing GSM/EDGE standards, includefast physical layer (L1) retransmission combining and link adaptationtechniques. These techniques may deliver significantly improved packetdata throughput performance between the mobile terminals 110 and 112 andthe BS 114.

The HSDPA technology employs several important new technologicaladvances. Some of these may comprise scheduling for the downlink packetdata operation at the BS 114, higher order modulation, adaptivemodulation and coding, hybrid automatic repeat request (HARQ), physicallayer feedback of the instantaneous channel condition, and a newtransport channel type known as high-speed downlink shared channel(HS-DSCH) that allows several users to share the air interface channel.When deployed, HSDPA may co-exist on the same carrier as the currentWCDMA and UMTS services, allowing operators to introduce greatercapacity and higher data speeds into existing WCDMA networks. HSDPAreplaces the basic features of WCDMA, such as variable spreading factorand fast power control, with adaptive modulation and coding, extensivemulticode operation, and fast and spectrally efficient retransmissionstrategies.

In current-generation WCDMA networks, power control dynamics are on theorder of 20 dB in the downlink and 70 dB in the uplink. WCDMA downlinkpower control dynamics are limited by potential interference betweenusers on parallel code channels and by the nature of WCDMA base stationimplementations. For WCDMA users close to the base station, powercontrol may not reduce power optimally, and reducing power beyond the 20dB may therefore have only a marginal impact on capacity. HSDPA, forexample, utilizes advanced link adaptation and adaptive modulation andcoding (AMC) to ensure all users enjoy the highest possible data rate.AMC therefore adapts the modulation scheme and coding to the quality ofthe appropriate radio link.

FIG. 1C illustrates an exemplary Layer 1 HARQ control situated in a basestation to remove retransmission-related scheduling and storing from theradio network controller, in connection with an embodiment of theinvention. Referring to FIG. 1C, there is shown a hybrid automaticrepeat request (HARQ) operation, which is an operation designed toreduce the delay and increase the efficiency of retransmissions. Layer 1HARQ control is situated in the Node B, or base station (BS), 122 thusremoving retransmission-related scheduling and storing from the radionetwork controller (RNC) 120. This HARQ approach avoids hub delay andmeasurably reduces the resulting retransmission delay.

For example, when a link error occurs, due to signal interference orother causes, a mobile terminal 124 may request the retransmission ofthe data packets. While current-generation WCDMA networks handle thoseretransmission requests through the radio network controller 120, HSDPAretransmission requests are managed at the base station 122.Furthermore, received packets are combined at the physical (PHY) layerand retrieved only if successfully decoded. If decoding has failed, thenew transmission is combined with the old transmission before channeldecoding. The HSDPA approach allows previously transmitted frames (thatfailed to be decoded) to be combined with the retransmission. Thiscombining strategy provides improved decoding efficiencies and diversitygains while minimizing the need for additional repeat requests.

While the spreading factor may be fixed, the coding rate may varybetween ¼ and ¾, and the HSDPA specification supports the use of up to10 multicodes. More robust coding, fast HARQ, and multi-code operationeliminates the need for variable spreading factor and also allows formore advanced receiver structures in the mobile such as equalizers asapposed to the traditional RAKE receiver used in most CDMA systems. Thisapproach may also allow users having good signal quality or highercoding rates and those at the more distant edge of the cell having lowercoding rates to each receive an optimum available data rate.

By moving data traffic scheduling to the base station 122, and thuscloser to the air interface, and by using information about channelquality, terminal capabilities, QoS, and power/code availability, HSDPAmay achieve more efficient scheduling of data packet transmissions.Moving these intelligent network operations to the base station 122allows the system to take full advantage of short-term variations, andthus to speed and simplify the critical transmission scheduling process.The HSDPA approach may, for example, manage scheduling to track the fastfading of the users and when conditions are favorable to allocate mostof the cell capacity to a single user for a very short period of time.At the base station 122, HSDPA gathers and utilizes estimates of thechannel quality of each active user. This feedback provides currentinformation on a wide range of channel physical layer conditions,including power control, ACK/NACK ratio, QoS, and HSDPA-specific userfeedback.

While WCDMA Release 99 or WCDMA Release 4 may support a downlink channel(DCH) or a downlink shared channel (DSCH), the HSDPA operation providedby WCDMA Release 5 may be carried on a high-speed downlink sharedchannel (HS-DSCH). This higher-speed approach uses a 2-ms interval framelength (also known as time transmit interval), compared to DSCH framelengths of 10, 20, 40 or 80 ms. DSCH utilizes a variable spreadingfactor of 4 to 256 chips while HS-DSCH may utilize a fixed spreadingfactor of 16 with a maximum of 15 codes. HS-DSCH may support 16-levelquadrature amplitude modulation (16-QAM), link adaptation, and thecombining of retransmissions at the physical layer with HARQ. HSDPA alsoleverages a high-speed shared control channel (HS-SCCH) to carry therequired modulation and retransmission information. An uplink high-speeddedicated physical control channel (HS-DPCCH) carries ARQacknowledgements, downlink quality feedback and other necessary controlinformation on the uplink.

FIG. 1D is a chart illustrating exemplary average carried loads forHSDPA-based macrocell and microcell systems, in connection with anembodiment of the invention. Referring to the chart 130 in FIG. 1D, inpractical deployments, HSDPA more than doubles the achievable peak userbit rates compared to WCDMA Release 99. With bit rates that arecomparable to DSL modem rates, HS-DSCH may deliver user bit rates 134 inlarge macrocell environments exceeding 1 Mbit/s, and rates 140 in smallmicrocells up to 5 Mbit/s. The HSDPA approach supports bothnon-real-time UMTS QoS classes and real-time UMTS QoS classes withguaranteed bit rates.

Cell throughput, defined as the total number of bits per secondtransmitted to users through a single cell, increases 100% with HSDPAwhen compared to the WCDMA Release 99. This is because HSDPA's use ofHARQ combines packet retransmission with the earlier transmission, andthus no transmissions are wasted. Higher order modulation schemes, suchas 16-QAM, enable higher bit rates than QPSK-only modulation in WCDMARelease 99, even when the same orthogonal codes are used in bothsystems. The highest throughput may be obtained with low inter-pathinterference and low inter-cell interference conditions. In microcelldesigns, for example, the HS-DSCH may support up to 5 Mbit/s per sectorper carrier, or 1 bit/s/Hz/cell.

FIG. 2A is a block diagram of an exemplary one-transmit (1-Tx) andmultiple-receive (M-Rx) antennas wireless communication system withreceiver channel weight generation, in accordance with an embodiment ofthe invention. Referring to FIG. 2A, a wireless communication system 200may comprise a single transmit antenna (Tx_1) 202, a first receiveantenna (Rx_1) 206, (M−1) additional receive antennas (Rx_2 208 to Rx_M209), (M−1) mixers 210 to 211, an adder 212, an RF processing block 214,a chip matched filter (CMF) 216, a cluster path processor (CPP) 218, abaseband (BB) processor 220, and a single weight baseband generator(SWBBG) 221. The SWBBG 221 may comprise a single weight generator (SWG)channel estimator 222 and an SWG algorithm block 224.

The transmit antenna, Tx_1 202, may comprise suitable hardware that maybe adapted to transmit SC communication signals, s_(T), from a wirelesstransmitter device. The first receive antenna, Rx_1 206, and the (M−1)additional receive antennas, Rx_2 208 to Rx_M 209, may comprise suitablehardware that may be adapted to receive in each at least a portion ofthe transmitted SC communication signals in a wireless receiver device.For example, the receive antenna Rx_1 206 may receive signal S_(R1), thereceive antenna Rx_2 208 may receive signal S_(R2), and the receiveantenna Rx_M 209 may receive signal S_(RM), where S_(R1) to S_(RM)correspond to the portion of the signal ST received by the receiveantennas Rx_1 206 to Rx_M 209 respectively. The propagation channelsthat correspond to the paths taken by the SC communication signalsreceived by the receive antennas Rx_1 206 to Rx_M 209 may be representedby h₁ to h_(M) respectively. In this regard, h₁ to h_(M) may representtime varying impulse responses of the RF paths taken by the portion ofthe transmitted SC communication signals received by the receiveantennas Rx_1 206 to Rx_M 209 respectively. In some instances, awireless transmitter device comprising a single transmit antenna may beadapted to periodically transmit calibration and/or pilot signals thatmay be utilized by an M-Rx antenna wireless receiver device to determineestimates of h₁ to h_(M). The 1-Tx M-Rx antennas wireless communicationsystem 200 in FIG. 2A may represent a single-input single-output (SIMO)communication system.

The mixers 210 to 211 may comprise suitable logic and/or circuitry thatmay be adapted to operate as a complex multiplier that may modify thephase of the portion of the SC communication signals received by thereceive antennas Rx_2 208 to Rx_M 209 via a rotation waveforms e^(jw)^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t), where w_(rk)=2πl f_(rk) and f_(rk)is the rotation frequency that preserves the orthogonality of thereceived signals at the multiple receiving antennas Rx_1 206 to Rx_M209. The rotation frequency that preserves the signal orthogonality atthe receiving antennas may be selected as f_(rk)=kf_(r) where k=1, 2, 3. . . M−1. Other rotation waveforms such as triangular or squarewaveforms may be utilized with the same frequency relationships. Inaddition, waveforms representing different orthogonal codes of the samefrequency may be utilized, similar to the CDMA orthogonal codes with thesame spreading. In this regard, the following exemplary sequences may beutilized: the first receive antenna Rx_1 206 may utilize the sequence [11 1 1], the second receive antenna Rx_2 208 may utilize the sequence [−1−1 1 1], a third receive antenna (Rx_3) may utilize the sequence [—1 1−1 1], and so on. In this embodiment, e^(jw) ^(rk) ^(t) is used as anexemplary waveform. In some implementations, the mixers 210 to 211 maycomprise a variable gain amplifier and a phase shifter, for example.

The channel weights comprising phase components for the rotationwaveforms may be provided by the SWBBG 221 for modifying the signalsreceived by the receive antennas Rx_2 208 to Rx_M 209 to achieve channelorthogonality between the receive antenna Rx_1 206 and the receiveantennas Rx_2 208 to Rx_M 209. Moreover, the channel weights may alsocomprise amplitude components that modify the received signals. In someinstances, the output of the mixers 210 to 211 may be transferred to abandpass filter and/or a low noise amplifier (LNA) for furtherprocessing of the received signals. The adder 212 may comprise suitablehardware, logic, and/or circuitry that may be adapted to add the outputof the receive antenna Rx_1 206 with the output of the mixers 210 to 211to generate a combined received SC communication signal, S_(RC), or gainbalanced point. In some instances, bringing the output signals of thereceive antenna Rx_1 206 and the mixers 210 to 211 together into asingle electrical connection may provide the functionality of the adder212. Notwithstanding, an output of the adder 212 may be transferred tothe RF block 214 for further processing of the combined received SCcommunication signal, s_(RC).

The RF block 214 may comprise suitable logic and/or circuitry that maybe adapted to process the combined received SC communication signal,S_(RC). The RF block 214 may perform, for example, filtering,amplification, and/or analog-to-digital (A/D) conversion operations. TheCMF 216 may comprise suitable logic, circuitry, and/or code that may beadapted to operate as a matched-filter on the digital output from the RFblock 214. The output of the CMF 216 may be transferred, for example, tothe CPP 218 and/or to the BB processor 220 for further processing. TheBB processor 220 may comprise suitable logic, circuitry, and/or codethat may be adapted to digitally process the filtered output of the CMF216 to determine an estimate of the transmitted SC communicationsignals, ŝ_(T).

The CPP 218 may comprise suitable logic, circuitry, and/or code that maybe adapted to process the filtered output of the CMF 216 to determine atleast a baseband combined channel estimate, which may compriseinformation regarding propagation channels h₁ to h_(M). For example, aportion of h may comprise information regarding the propagation channelsbetween the transmit antenna Tx_1 202 and the receive antennas Rx_1 206and Rx_2 208, that is, h₁ and h₂, while another portion of h maycomprise information regarding the propagation channel between thetransmit antenna Tx_1 202 and the receive antenna Rx_M 209, that is,h_(M). The channel estimation filter utilized in the CPP 218 may beselected such that the channel estimates may track the changes in therotation waveforms applied to the mixers 210 through 211. The CPP 218may also be adapted to generate a lock indicator signal that may beutilized by, for example, the BB processor 220 as an indication ofwhether the channel estimates are valid. The CPP 118 may be adapted toprocess the received signals in clusters. U.S. application Ser. No.11/173,854 provides a detailed description of signal clusters and ishereby incorporated herein by reference in its entirety.

The SWBBG 221 may comprise suitable logic, circuitry, and/or code thatmay be adapted to receive the baseband combined channel estimate, ĥ,from the CPP 218 and determine (M−1) phase components for the rotationwaveforms e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t) to be applied bymixers 210 to 211 to modify the portion of the SC communication signalsreceived by the receive antennas Rx_2 208 to Rx_M 209. The SWG channelestimator 222 may comprise suitable logic, circuitry, and/or code thatmay be adapted to process the baseband combined channel estimate, ĥ,generated by the CPP 218 and determine a vector Ĥ_(M×1) of propagationchannel estimates ĥ₁ to ĥ_(M), where ĥ₁ is an estimate of the timevarying impulse response h₁ and ĥ_(M) is an estimate of the time varyingimpulse response h_(M). The actual time varying impulse responses, h₁ toh_(M), may contain multiple propagation paths arriving at differentdelays. The SWG algorithm block 224 may comprise suitable logic,circuitry, and/or code that may be adapted to utilize the contents ofthe vector Ĥ_(M×1) to determine (M−1) channel weights that may compriseamplitude and phase components, A₁ to A_(M−1) and φ₁ to φ_(M−1).

FIG. 2B is a block diagram of an exemplary two-transmit (2-Tx) andmultiple-receive (M-Rx) antennas wireless communication system withreceiver channel weight generation, in accordance with an embodiment ofthe invention. Referring to FIG. 2B, the wireless communication system250 may comprise a dedicated physical channel (DPCH) block 226, aplurality of mixers 228, 230 and 232, a first combiner 234, a secondcombiner 236, a first transmit antenna (Tx_1) 238, and an additionaltransmit antenna (Tx_2) 240. The receiver portion of the wirelesscommunication system 250 may be similar to that of the wirelesscommunication system 200 in FIG. 2A and may comprise a first receiveantenna (Rx_1) 206, (M−1) additional receive antennas (Rx_2 208 to Rx_M209), (M−1) mixers 210 to 211, an adder 212, an RF block 214, a CMF 216,a CPP 218, a BB processor 220, and a SWBBG 221. The SWBBG 221 maycomprise a SWG channel estimator 222 and a SWG algorithm block 224.

The DPCH 226 may comprise suitable logic, circuitry, and/or code thatmay be adapted to receive a plurality of input channels, for example, adedicated physical control channel (DPCCH) and a dedicated physical datachannel (DPDCH). The DPCH 226 may be adapted to simultaneously controlthe power on each of the DPCCH and DPDCH channels. The mixer 228 maycomprise suitable logic and/or circuitry that may be adapted to multiplythe output of DPCH 226 with a spread and/or scramble signal to generatea spread complex-valued signal that may be transferred to the inputs ofthe mixers 230 and 232.

The mixers 230 and 232 may comprise suitable logic and/or circuitry thatmay be adapted to multiply the spread complex-valued signal from themixer 228 with the closed loop 1 (CL1) and closed loop 2 (CL2) transmitdiversity weight factors W₁ and W₂ respectively. Closed loop transmitdiversity may be described in the 3^(rd) Generation Project Partnership(3GPP), Technical Specification Group Radio Access Network, PhysicalLayer Procedures (FDD), Release 6 (3GPP TS 25.214 V5.5.0, 2003-06). Forexample, the weight factors W₁ and W₂ may correspond to phase and/oramplitude component feedback adjustments that may be generated by thereceiver based on the type of space-time coding that is used. The SWBBG221 may be adapted to generate the weight factors W₁ and W₂. This may beapplied to closed loop transmit diversity as currently being used inWCDMA. In this regard, a closed loop processing block may be utilized totransfer the weight factors or parameters that correspond to thoseweight factors to the transmitter via an uplink feedback process. Theweight factors W₁, W₂ and the (M−1) channel weights at the receiveantennas Rx_2 through Rx_M may be determined by a joint computation bythe SWG algorithm 224 in the SWBBG 221. The (M−1) channel weights maycomprise amplitude and phase components, A₁ to A_(M−1) and φ₁ toφ_(M−1), respectively.

The output of the mixer 230 may be transferred to the first combiner 234and the output of the mixer 232 may be transferred to the secondcombiner 236. The first and second combiners 234 and 236 may comprisesuitable logic, circuitry, and/or code that may be adapted to add orcombine the outputs generated by mixers 230 and 232 with a common pilotchannel 1 (CPICH1) signal and a common pilot channel 2 (CPICH2) signalrespectively. The CPICH1 signal and CPICH2 signals may comprise fixedchannelization code allocation and may be utilized to measure the signalphase and amplitude and strength of the propagation channels between thetransmit antennas and the receive antennas.

The first transmit antenna, Tx_1 238, and the additional or secondtransmit antenna, Tx_2 240, may comprise suitable hardware that may beadapted to transmit a plurality of SC communication signals, ST, from awireless transmitter device. The propagation channels that correspond tothe paths taken by the SC communication signals transmitted from thetransmit antennas Tx_1 238 and Tx_2 240 and received by the receiveantennas Rx_1 206 to Rx_M 209 may be represented by an M×2 matrix,H_(M×2). The matrix H_(M×2) may comprise propagation channels h₁₁ toh_(M1) and h₁₂ to h_(M2). In this regard, h₁₁ to h_(M1) may representthe time varying impulse responses of the RF paths taken by the portionof the transmitted SC communication signals transmitted by transmitantenna Tx_1 238 and received by the receive antennas Rx_1 206 to Rx_M209 respectively. Similarly, h₁₂ to h_(M2) may represent the timevarying impulse responses of the RF paths taken by the portion of thetransmitted SC communication signals transmitted by transmit antennaTx_2 240 and received by the receive antennas Rx_1 206 to Rx_M 209respectively. In some instances, a wireless transmitter devicecomprising a first and a second transmit antenna may be adapted toperiodically transmit calibration and/or pilot signals that may beutilized by an M-Rx antenna wireless receiver device to determineestimates of h₁₁ to h_(M1) and h₁₂ to h_(M2). The 2-Tx and M-Rx antennaswireless communication system 250 in FIG. 2B may represent a MIMOcommunication system whereby the diversity gain may be increased for thetransmitted data.

The CPP 218 in FIG. 2B may be adapted to determine a first basebandcombined channel estimate, ĥ₁, which may comprise information regardingpropagation channels h₁₁ to h_(M1). For example, a portion of ĥ₁ maycomprise information regarding the propagation channels between thetransmit antenna Tx_1 238 and the receive antennas Rx_1 206 and Rx_2208, that is, h₁₁ and h₂₁, while another portion of ĥ₁ may compriseinformation regarding the propagation channels between the transmitantenna Tx_1 238 and the receive antennas Rx_1 206 and Rx_M 209, thatis, ĥ₁₁ and h_(M1). The actual time varying impulse responses h₁₁ toh_(M1) may comprise multiple propagation paths arriving at differenttime delays.

The CPP 218 in FIG. 2B may also be adapted to determine a secondbaseband combined channel estimate, ĥ₂, which may comprise informationregarding propagation channels h₁₂ to h_(M2). For example, a portion ofĥ₂ may comprise information regarding the propagation channels betweenthe transmit antenna Tx_2 240 and the receive antennas Rx_1 206 and Rx_2208, that is, h₁₂ and h₂₂, while another portion of ĥ₂ may compriseinformation regarding the propagation channels between the transmitantenna Tx_2 240 and the receive antennas Rx_1 206 and Rx_M 209, thatis, h₁₂ and h_(M2). The actual time varying impulse responses h₁₂ toh_(M2) may comprise multiple propagation paths arriving at differenttime delays. The combined channel estimates may be determined, that is,may be separated, in the CPP 218 utilizing the orthogonal relationshipbetween the common pilot signals CPICH1 and CPICH2 that may betransmitted by the antennas Tx_1 238 and Tx_2 240, respectively.

The SWG channel estimator 222 in FIG. 2B may be adapted to process thefirst and second baseband combined channel estimates, ĥ₁ and ĥ₂,determined by the CPP 218 and may determine a matrix Ĥ_(M×2) ofpropagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₂ to ĥ_(M2), whichcorrespond to estimates of the matrix H_(M×2) of time varying impulseresponses h₁₁ to h_(M1) and h₁₂ to h_(M2), respectively. The SWGalgorithm block 224 may utilize the contents of the matrix Ĥ_(M×2) todetermine (M−1) channel weights to be applied to the mixers 210 to 211to modify the portions of the transmitted SC communication signalsreceived by the additional receive antennas Rx_2 208 to Rx_M 209 so thatthe receiver SINR is maximized, for example. The (M−1) channel weightsmay comprise amplitude and phase components, A₁ to A_(M−1), and φ₁ toφ_(M−1), for example. Moreover, the SWG algorithm block 224 may beadapted to generate the weight factors W₁ and W₂ joint or concurrentlywith the (M−1) channel weights.

FIG. 3A is a flow diagram illustrating exemplary steps for channelestimation in a 2-Tx and M-Rx antennas wireless communication system, inaccordance with an embodiment of the invention. Referring to FIG. 3A,after start step 302, in step 304, the SC communication signals, ST, maybe transmitted from the transmit antennas Tx_(—)1 238 and Tx_2 240 inFIG. 2B. In step 306, the first and additional receive antennas, Rx_1206 to Rx_M 209, may receive a portion of the transmitted SCcommunication signals. In step 308, the signals received by theadditional receive antennas Rx_1 206 to Rx_M 209 may be multiplied by,for example, rotation waveforms, such as sine, square, or triangularwaveforms for example, in the mixers 210 to 211 in FIG. 2B. In thisregard, the rotation waveforms may have a given set of phase componentvalues. In step 310, the output of the receive antenna Rx_1 206 and theoutput of the mixers 210 to 211 associated with the additional receiveantennas Rx_2 208 to Rx_M 209 may be added or combined into the receivedSC communication signal, S_(RC). The combination may occur in the adder212, for example.

In step 312, the CPP 218 may determine the first and second basebandcombined channel estimates, ĥ₁ and ĥ₂, which comprise informationregarding propagation channels h₁₁ to h_(M1) and h₁₂ to h_(M2). In step314, the SWG channel estimator 222 in the SWBBG 221 may determine thematrix Ĥ_(M×2) of propagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ toĥ_(M2). In this regard, the propagation channel estimates ĥ₁₁ to ĥ_(M1)and ĥ₁₂ to ĥ_(M2) may be determined concurrently.

In step 316, the (M−1) maximum SINR channel weights that compriseamplitude and phase components, A₁ to A_(M−1) and φ₁ to φ_(M−1), may begenerated concurrently. The weight factors W₁ and/or W₂ in FIG. 2B maybe generated concurrently with the (M−1) maximum SINR channel weights.In some instances, the channel weights may be based on the propagationchannel estimates determined after the weight factors W₁ and W₂ to thetransmitter. In step 318, additional SC communication signals receivedmay be phase and amplitude adjusted based on the maximum SINR channelweights applied to the mixers 210 to 211. The channel estimation phaserotation and the maximum SINR phase/amplitude adjustment described inflow chart 300 may be performed continuously or may be performedperiodically. In this regard, FIG. 3B illustrates an exemplary periodicphase rotation for an in-phase (I) signal received in one of theadditional receive antennas, in accordance with an embodiment of theinvention.

FIG. 4A is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and 2-Rx antennassystem, in accordance with an embodiment of the invention. Referring toFIG. 4A, a receiver system 400 may comprise a first receive antenna(Rx_1) 402, an additional receive antenna (Rx_2) 404, an adder 406, amixer 408, and a single weight baseband generator (SWBBG) 410. The SWBBG410 may comprise a phase rotator start controller 414, a delay block416, a single weight generator (SWG) channel estimator 418, an SWGalgorithm block 420, and an RF phase and amplitude controller 412. TheSWBBG 410 may represent an exemplary implementation of the SWBBG 221 inFIG. 2B.

The first receive antenna, Rx_1 402, and the additional or secondreceive antenna, Rx_2 404, may comprise suitable hardware that may beadapted to receive at least a portion of transmitted SC communicationsignals in the receiver system 400. For example, the receive antennaRx_1 402 may receive a signal S_(R1) while the receive antenna Rx_2 404may receive a signal S_(R2). The mixer 408 may correspond to, forexample, the mixer 210 in FIG. 2B. In some instances, the output of themixer 308 may be communicated to a bandpass filter and/or a low noiseamplifier (LNA) for further processing of the received signals.

The adder 406 may comprise suitable hardware, logic, and/or circuitrythat may be adapted to add the output of the receive antenna Rx_1 402and the output of the mixer 408 to generate a combined received SCcommunication signal, S_(RC). In some instances, bringing the outputsignals of the receive antenna Rx_1 402 and the mixer 408 together intoa single electrical connection may provide the functionality of theadder 406. The output of the adder 406 may be transferred to additionalprocessing blocks for RF and baseband processing of the combinedreceived SC communication signal, S_(RC).

The phase rotator and start controller 414 may comprise suitable logic,circuitry, and/or code that may be adapted to control portions of theoperation of the RF phase and amplitude controller 412 and to controlthe delay block 416. The phase rotator and start controller 414 mayreceive a signal, such as a reset signal, from, for example, the BBprocessor 220 in FIG. 2B, or from firmware operating in a processor, toindicate the start of operations that determine the propagation channelestimates and/or the channel weight to apply to the mixer 408. The delayblock 416 may comprise suitable logic, circuitry, and/or code that maybe adapted to provide a time delay to compensate for the RF/modem delay.The delay may be applied in order to compensate for the interval of timethat may occur between receiving the combined channel estimates, ĥ₁ andĥ₂, modified by the rotation waveform and the actual rotating waveformat the mixer 408.

The SWG channel estimator 418 may comprise suitable logic, circuitry,and/or code that may be adapted to process the first and second basebandcombined channel estimates, ĥ₁ and ĥ₂, and determine the matrix H_(2×2)of propagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂. The SWG channelestimator 418 may also be adapted to generate an algorithm start signalto the SWG algorithm block 420 to indicate that the propagation channelestimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂ are available for processing. In thisregard, the algorithm start signal may be asserted when integrationoperations performed by the SWG channel estimator 418 have completed.

The SWG algorithm block 420 may comprise suitable logic, circuitry,and/or code that may be adapted to determine a channel weight to betransferred to the mixer 408 via the RF phase and amplitude controller412 to modify the signal S_(R2). The channel weight to be transferred tothe mixer 408 may refer to the phase, φ, and amplitude, A. The channelweight may be based on the propagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁,and ĥ₂₂ and on additional information such as noise power estimates andinterference propagation channel estimates, for example. The SWGalgorithm block 420 may also be adapted to generate an algorithm endsignal to indicate to the RF phase and amplitude controller 412 that thechannel weight has been determined and that it may be applied to themixer 408. The SWG algorithm block 420 in FIG. 4A may also be adapted todetermine the weight factors W₁ and W₂. The channel weights and theweight factors W₁ and W₂ may be calculated jointly to maximize thereceiver SINR, for example.

The RF phase and amplitude controller 412 may comprise suitable logic,circuitry, and/or code that may be adapted to apply the rotationwaveform e^(jw) ^(r) ^(t) to the mixer 408. When phase and amplitudecomponents, A and φ, that correspond to the channel weight determined bythe SWG algorithm block 420 are available, the RF phase and amplitudecontroller 412 may apply amplitude A and phase φ to the mixer 408. Inthis regard, the RF phase and amplitude controller 412 may apply therotation waveform or the amplitude and phase components in accordancewith the control signals provided by the phase rotator start controller414 and/or the algorithm end signal generated by the SWG algorithm block420.

The phase rotation operation performed on the S_(R2) signal in theadditional receive antenna Rx_2 404 may be continuous or periodic. Acontinuous rotation of the S_(R2) signal may be perceived by a wirelessmodem as a high Doppler, and for some modem implementations this maydecrease the modem's performance. When a periodic rotation operation isutilized instead, the period between consecutive phase rotations maydepend on the Doppler frequency perceived by the wireless modem. Forexample, in a higher Doppler operation, it may be necessary to performmore frequent channel estimation while in a lower Doppler operation,channel estimation may be less frequent. The signal rotation period mayalso depend on the desired wireless modem performance and the accuracyof the propagation channel estimation. For example, when the Dopplerfrequency is 5 Hz, the period between consecutive rotations may be 1/50sec., that is, 10 rotations or channel estimations per signal fade.

FIG. 4B is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and M-Rx antennassystem, in accordance with an embodiment of the invention. Referring toFIG. 4B, a receiver system 430 may differ from the receiver system 400in FIG. 4A in that (M−1) additional receive antennas, Rx_2 404 to Rx_M405, and (M−1) mixers 408 to 409 may be utilized. In this regard, theSWG channel estimator 418 may be adapted to process the first and secondbaseband combined channel estimates, ĥ₁ and h₂, and determine the matrixH_(M×2) of propagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ toĥ_(M2).

The SWG algorithm block 420 may also be adapted to determine (M−1)channel weights, that may be utilized to maximize receiver SINR, forexample, to be applied to the mixers 408 to 409 to modify the portionsof the transmitted SC communication signals received by the additionalreceive antennas Rx_2 404 to Rx_M 405. The (M−1) channel weights maycomprise amplitude and phase components, A₁ to A_(M−1) and φ₁ toφ_(M−1). The SWG algorithm block 420 in FIG. 4B may also be adapted todetermine the weight factors W₁ and W₂ that may be applied to the mixers230 and 232 in FIG. 2B. The channel weights and the weight factors W₁and W₂ may be calculated jointly to maximize the receiver SINR, forexample.

The RF phase and amplitude controller 412 may also be adapted to applyrotation waveforms e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t) or phaseand amplitude components, A₁ to A_(M−1) and φ₁ to φ_(M−1), to the mixers408 to 409. In this regard, the RF phase and amplitude controller 312may apply the rotation waveforms or the amplitude and phase componentsin accordance with the control signals provided by the phase rotatorstart controller 414 and/or the algorithm end signal generated by theSWG algorithm block 420.

FIG. 4C is a block diagram of an exemplary RF phase and amplitudecontroller, in accordance with an embodiment of the invention. Referringto FIG. 4C, the RF phase and amplitude controller 412 may comprise aswitch 440, a plurality of rotation waveform sources 442, and aplurality of SWG algorithm weights 444. The switch 440 may comprisesuitable hardware, logic, and/or circuitry that may be adapted to selectbetween the rotation waveforms e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1))^(t) and the SWG algorithm determined weights A₁e^(jφ) ¹ toA_(M−1)e^(jφ) ^(M−1) . The rotation waveform sources 442 may comprisesuitable hardware, logic and/or circuitry that may be adapted togenerate the signal e^(jw) ^(rk) ^(t), where w_(rk)=2πf_(rk) and f_(rk)is the rotation frequency that preserves the orthogonality of thereceived signals at the receive antennas Rx_2 402 to Rx_M 405 in FIG.4B, for example. The rotation frequency that preserves the signalorthogonality at the receiving antennas may be selected as w_(rk)=kw_(r)where k=1, 2, . . . , M−1. Other rotation waveforms such as triangularor square waveforms may be utilized with the same frequencyrelationships. Moreover, waveforms representing different orthogonalcodes of the same frequency may also be utilized, similar to the CDMAorthogonal codes with the same spreading. In this embodiment, the signale^(jw) ^(rk) ^(t) may be utilized as an exemplary waveform. Theplurality of SWG algorithm weights 344 may comprise suitable hardware,logic, and/or circuitry that may be adapted to generate the signalsA₁e^(jφ) ¹ to A_(M−1)e^(jφ) ^(M−1) from the amplitude and phasecomponents, A₁ to A_(M−1) and φ₁ to φ_(M−1), respectively.

In operation, the RF phase and amplitude controller 412 may apply thesignals e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t) to the mixers 408 to409 in FIG. 4B based on control information provided by the phaserotator start controller 414. The switch 440 may select the rotationwaveform sources 442 based on the control information provided by thephase rotator start controller 414. Once the channel weights aredetermined by the SWG algorithm block 420 and the phase and amplitudecomponents have been transferred to the RF phase and amplitudecontroller 412, the algorithm end signal may be utilized to change theselection of the switch 440. In this regard, the switch 440 may beutilized to select and apply the signals A₁e^(jφ) ¹ to A_(M−1)e^(jφ)^(M−1) to the mixers 408 to 409 in FIG. 4B.

FIG. 5 is a flow diagram illustrating exemplary steps in the operationof the single weight baseband generator (SWBBG) that may be utilized forchannel weight generation in a 2-Tx and M-Rx antennas system, inaccordance with an embodiment of the invention. Referring to FIG. 5,after start step 502, in step 504, the phase rotator start controller414 in FIG. 4B may receive the reset signal to initiate operations fordetermining propagation channel estimates and channel weights in theSWBBG 410. The phase rotator start controller 414 may generate controlsignals to the delay block 416 and to the RF phase and amplitudecontroller 412. The control signals to the delay block 416 may beutilized to determine a delay time to be applied by the delay block 416.The control signals to the RF phase and amplitude controller 412 may beutilized to determine when to apply the rotation waveforms that havebeen modified by the channel weights to the mixers 408 to 409.

In step 506, the RF phase and amplitude controller 412 may applyrotation waveforms, such as those provided by the rotation waveformsources 442 in FIG. 4C, to the mixers 408 to 409 in FIG. 4B. In step508, the delay block 416 may apply a time delay signal to the SWGchannel estimator 418 to reflect the interval of time that may occurbetween receiving the SC communication signals and when the first andsecond baseband combined channel estimates, ĥ₁ and ĥ₂, are available tothe SWG channel estimator 418. For example, the time delay signal may beutilized as an enable signal to the SWG channel estimator 418, where theassertion of the time delay signal initiates operations for determiningpropagation channel estimates. In step 510, the SWG channel estimator418 may process the first and second baseband combined channelestimates, ĥ₁ and ĥ₂, and may determine the matrix Ĥ_(M×2) ofpropagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ to ĥ_(M2). The SWGchannel estimator 418 may transfer the propagation channel estimates ĥ₁₁to ĥ_(M1) and ĥ₁₂ to ĥ_(M2) to the SWG algorithm block 420. In step 512,the SWG channel estimator 418 may generate the algorithm start signaland may assert the signal to indicate to the SWG algorithm block 420that it may initiate operations for determining channel weights.

In step 514, the SWG algorithm block 420 may determine the channelweights comprising phase and amplitude components, A₁ to A_(M−1) and φ₁to φ_(M−1), based on the propagation channel estimates ĥ₁₁ to ĥ_(M1) andĥ₁₂ to ĥ_(M2) and/or noise power estimates, for example. The SWGalgorithm block 420 may transfer the channel weights to the RF phase andamplitude controller 412. In some instances, the SWG algorithm block 420may also generate the weight factors W₁ and/or W₂. In step 516, the SWGalgorithm block 420 may generate the algorithm end signal to indicate tothe RF phase and amplitude controller 412 that the channel weights areavailable to be applied to the mixers 408 to 409. In step 518, the RFphase and amplitude controller 412 may apply the rotation waveforms withphase and amplitude components, A₁ to A_(M−1) and φ₁ to φ_(M−1), to themixers 408 to 409, in accordance with the control signals provided bythe phase rotator start controller 414.

In step 520, the receiver system 430 in FIG. 4B may determine whetherthe phase rotation operation on the received SC communication signals isperiodic. When the phase rotation operation is not periodic butcontinuous, the process may proceed to step 508 where a delay may beapplied to the SWG channel estimator 418. In instances when the phaserotation operation is periodic, the process may proceed to step 522where the receiver system 430 may wait until the next phase rotationoperation is initiated by the reset signal. In this regard, the processcontrol may proceed to step 504 upon assertion of the reset signal tothe phase rotator start controller 414.

FIG. 6 is a flow diagram illustrating exemplary steps for determiningchannel weights in additional receive antennas utilizing signal-to-noiseratio (SNR) or signal-to-interference-and-noise ratio (SINR), inaccordance with an embodiment of the invention. Referring to FIG. 6,after start step 602, in step 604, the SWG algorithm block 420 maydetermine whether the signals received in the receive antennas are noiselimited. The SWG algorithm block 420 may receive noise statistics and/orother noise information from either the CPP 218 and/or from the BBprocessor 220. When the received signals are noise limited, the flowdiagram control may proceed to step 608. In step 608, the SWG algorithmblock 420 may generate models for the received signals. For example, themodels for a 1-Tx and 2-Rx antennas system may be represented by thefollowing expressions:r ₁ =h ₁ s+n ₁,r ₂ =Ae ^(jθ) h ₂ s+Ae ^(jθ) n ₂, andy=r ₁ +r ₂ =s(h ₁ +Ae ^(jθ) h ₂)+n ₁ +Ae ^(jθ) n ₂,where r₁ may represent a model of the signal received in a first receiveantenna, r₂ may represent a model of the signal received in the secondreceive antenna, s may represent the transmitted signal, and n₁ mayrepresent a noise component at the first receive antenna, whose timevarying impulse response is represented by h₁. The parameter n₂ mayrepresent a noise component at the second receive antenna, whose timevarying impulse response is represented by h₂, θ may represent the phasefactor between the signal received in the first and second receiveantennas, and A may represent an amplitude factor. The parameter y mayrepresent the sum of the received signal models and may comprise acombined signal component s(h₁+Ae^(jθ)h₂) and a combined noise componentn₁+Ae^(jθ)n₂.

In step 610, the received signal models may be utilized to determine asignal strength parameter. In this regard, the signal-to-noise ratio(SNR) may correspond to the signal strength parameter to be determined.For example, for a 1-Tx and 2-Rx antennas system, the SNR may bedetermined by maximizing the following expression for various phase, θ,and amplitude, A, factors:

${SNR} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\vartheta}h_{2}}}}^{2}}{{E{n_{1}}^{2}} + {E{{A\;{\mathbb{e}}^{j\vartheta}n_{2}}}^{2}}} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\vartheta}h_{2}}}}^{2}}{\sigma^{2}\left( {1 + A^{2}} \right)}.}}$The SNR numerator may correspond to the y parameter's combined signalcomponent while the SNR denominator may correspond to the y parameter'scombined noise component. The phase factor, θ, may be selected, forexample, from a 360-degrees phase rotation while the amplitude factor,A, may be selected, for example, from an set amplitude range. In oneembodiment of the invention, the phase factor may be varied in aplurality of phase factor steps over the 360-degrees phase rotation tofind the maximum SNR value. In another embodiment of the invention, thephase factor may be varied in a plurality of phase factors steps overthe 360-degrees phase rotation and the amplitude factor may be varied ina plurality of amplitude factor values over the amplitude range to findthe maximum SNR value.

In step 620, after determining the maximum SNR in step 610, the SWGalgorithm block 420 may utilize the amplitude factor and phase factorthat corresponds to the maximum SNR to determine the amplitude and phaseto be provided to the RF amplitude and phase controller 412 in step 620.For example, in one embodiment of the invention, the amplitude and/orphase factors that correspond to the maximum SNR may be utilized as theamplitude and phase to be transferred to the RF amplitude and phasecontroller 412. After application of the appropriate amplitude and phaseby the RF amplitude and phase controller 412 to the receive antennas,the flow diagram control may proceed to end step 622 until a next phaseand amplitude determination is necessary.

Returning to step 604, when received signals are not noise limited, theflow control may proceed to step 606 where a determination may be madeas to whether multiple interfering signals may be present and may needto be considered during channel weight determination. When a singleinterferer is considered, the flow diagram control may proceed to step612. In step 612 the SWG algorithm block 420 may generate models for thereceived signals. For example, the models for a 1-Tx and 2-Rx antennassystem may be represented by the following expressions:r ₁ =h ₁ s+h _(l1) s _(l) +n ₁,r ₂ =Ae ^(jθ)(h ₂ s+h _(l2) s _(l) +n ₂), andy=r ₁ +r ₂ =s(h ₁ +Ae ^(jθ) h ₂)+n ₁ +s _(l)(h _(l1) +Ae ^(jθ) h_(l2))+Ae ^(jθ) n ₂,where r₁ may represent a model of the signal received in a first receiveantenna, r₂ may represent a model of the signal received in the secondreceive antenna, s may represent the transmitted signal, s_(l) mayrepresent the interference signal, and n₁ may represent a noisecomponent at the first receive antenna whose time varying impulseresponse is h₁. The parameter n₂ may represent a noise component at thesecond receive antenna whose time varying impulse response is h₂, θ mayrepresent the phase factor between the signal received in the first andsecond receive antennas, and A may represent an amplitude factor.Moreover, the time varying impulse response h_(l1) may correspond to thepropagation channel between the interference signal source and the firstreceive antenna and the time varying impulse response h_(l2) maycorrespond to the propagation channel between the interference signalsource and the second receive antenna. The parameter y may represent thesum of the received signal models and may comprise a combined signalcomponent s(h₁+Ae^(jθ)h₂) and a combined noise plus interferencecomponent n₁+s_(l)(h_(l1)+Ae^(jθ)h_(l2))+Ae^(jθ)n₂.

In step 614, the received signal models may be utilized to determine asignal strength parameter. In this regard, thesignal-to-interference-and-noise ratio (SINR) may correspond to thesignal strength parameter to be determined. For example, for a 1-Tx and2-Rx antennas system, the SINR may be determined by maximizing thefollowing expression for various phase, θ, and amplitude, A, factors:

${SINR} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\vartheta}h_{2}}}}^{2}}{{E{n_{1}}^{2}} + {E{{A\;{\mathbb{e}}^{j\vartheta}n_{2}}}^{2}} + {{h_{l\; 1} + {A\;{\mathbb{e}}^{j\vartheta}h_{l\; 2}}}}^{2}} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\vartheta}h_{2}}}}^{2}}{{\sigma^{2}\left( {1 + A^{2}} \right)} + {{h_{l\; 1} + {A\;{\mathbb{e}}^{j\vartheta}h_{l\; 2}}}}^{2}}.}}$where σ² is the noise power. The above SINR equations may be easilyextended to the SC MIMO case. The transmit antennas may include CL1 orCL2 transmit diversity weights. The joint transmit-received solution maybe formed in that case that may include the transmit CL weights and theadditional transmit antenna channel components in the SINR numerator.The SINR numerator may correspond to the y parameter's combined signalcomponent while the SINR denominator may correspond to the y parameter'scombined noise plus interference component. The phase factor, θ, may beselected, for example, from a 360-degrees phase rotation while theamplitude factor, A, may be selected, for example, from an set amplituderange. In one embodiment of the invention, the phase factor may bevaried in a plurality of phase factor steps over the 360-degrees phaserotation to find the maximum SNR value. In another embodiment of theinvention, the phase factor may be varied in a plurality of phasefactors steps over the 360-degrees phase rotation and the amplitudefactor may be varied in a plurality of amplitude factor values over arange of amplitudes to find the maximum SINR value.

After determining the SINR in step 614, the SWG algorithm block 420 maydetermine the amplitude and phase to be provided to the RF amplitude andphase controller 412 in step 620. After application of the appropriateamplitude and phase by the RF amplitude and phase controller 412, theflow diagram control may proceed to end step 622 until a next phase andamplitude determination is necessary.

After determining the SINR in step 618, the SWG algorithm block 420 maydetermine the amplitude and phase to be provided to the RF amplitude andphase controller 412 in step 620. After application of the appropriateamplitude and phase by the RF amplitude and phase controller 412, theflow diagram control may proceed to end step 622 until a next phase andamplitude determination is necessary.

The operations to maximize the signal strength described for steps 610,614, and 618 may be based on a search algorithm. In an exemplaryembodiment of the invention, a search algorithm may be utilized tosearch over 360-degrees phase rotation in 45-degree or 90-degree phasefactor steps and over a 0-5 amplitude range in 0.25 amplitude values orsteps, for example. For a 1-Tx and 2-Rx antenna system, with 90-degreephase factor steps, a phase only search algorithm may calculate 4 SNR orSINR values, for example. For a 2-Tx and 2-Rx antenna system with spacetime transmit diversity (STTD) transmit mode, with 90-degree phasefactor steps, a phase only search algorithm may calculate 4 SNR or SINRvalues. The maximum value generated by the algorithm may be the outputof the search algorithm. The SNR for the 2-Tx and 2-Rx antenna systemwith STTD transmit mode, for example, may be obtained as follows:

h₁ = h₁₁ + h₁₂ h₂ = h₂₁A 𝕖^(jϕ) + h₁₂A 𝕖^(jϕ), h = h₁₁ + h₁₂ + h₂₁A 𝕖^(jϕ) + h₁₂A 𝕖^(jϕ), and${SNR} = {\frac{{{h_{11} + h_{12} + {A\;{{\mathbb{e}}^{j\phi}\left( {h_{21} + h_{22}} \right)}}}}^{2}}{\sigma^{2}\left( {1 + A^{2}} \right)}.}$

For a 2-Tx and 2-Rx antenna system with the CL1 diversity mode, with90-degree phase factor steps at both receiver and transmitter, a phaseonly search algorithm may calculate 4×4=16 SNR or SINR values. In CL1diversity mode, the weight factor W₁ may be set to W₁=1, for example.For a 2-Tx and 2-Rx antenna system with the CL2 diversity mode, with90-degree phase factor steps at the receiver and 45-degree phase factorsteps and two power scaling weight levels at the transmitter, a phaseonly search algorithm may calculate 4×8×2=64 SNR or SINR values, forexample. The maximum value generated by the algorithm may be the outputof the search algorithm. The SNR for the 2-Tx and 2-Rx antenna systemwith CL1 and CL2 modes, for example, may be obtained as follows:

   h₁ = A₁𝕖^(jϕ₁)h₁₁ + A₂𝕖^(jϕ₂)h₁₂,   h₂ = A 𝕖^(jϕ)(A₁𝕖^(jϕ₁)h₂₁ + A₂𝕖^(jϕ₂)h₂₂),   h = A₁𝕖^(jϕ₁)h₁₁ + A₂𝕖^(jϕ₂)h₁₂ + A 𝕖^(jϕ)(A₁𝕖^(jϕ₁)h₂₁ + A₂𝕖^(jϕ₂)h₂₂), and${SNR} = {\frac{{{{A_{1}{\mathbb{e}}^{{j\phi}_{1}}h_{11}} + {A_{2}{\mathbb{e}}^{{j\phi}_{2}}h_{12}} + {A\;{{\mathbb{e}}^{j\phi}\left( {{A_{1}{\mathbb{e}}^{{j\phi}_{1}}h_{21}} + {A_{2}{\mathbb{e}}^{{j\phi}_{2}}h_{22}}} \right)}}}}^{2}}{\sigma^{2}\left( {1 + A^{2}} \right)}.}$

In another embodiment of the invention, a closed-form mathematicalexpression may also be utilized to maximize the SNR and/or the SINR.Utilizing an algorithm or closed-form expression that maximizes the SINRor SNR may provide a good compromise between implementation complexityand performance gains. Notwithstanding, the invention is not limited inthis regard, and other channel weight algorithms may also be utilized.

FIG. 7 is a flow diagram illustrating exemplary steps for determiningthe channel weight for the 1-Tx (or 2-Tx) and 2-Rx antenna system bymonitoring phase rotation, in accordance with an embodiment of theinvention. Referring to FIG. 7, in another embodiment of the invention,the channel weights may be determined by the method described in flowdiagram 700. In step 704, after start step 702, the SWBBG 410 in FIG. 4Amay generate phase rotation components to be applied by the mixer 408 tothe additional receive antenna. Signals received by the second receiveantenna Rx_2 404 may be modified via rotation waveform e^(jw) ^(r) ^(t).In step 706, the SWBBG 410 may monitor the baseband combined channelestimates determined by the CPP 218 during the rotation operation. Thecombined channel estimates represent a sum of the individual propagationchannels at the two receiving antennas Rx_1 402 and Rx_2 404, modifiedby the rotation operation at the Rx_2 404 antenna. In the case of themultipath propagation channel components arriving at the differentdelays, the combined channel estimates may represent a sum of themagnitudes square of the individual combined multipath components. Forexample, the rotation operation may occur over a 360-degrees phasefactor rotation.

The baseband combined channel estimates may be proportional to the powerof the received signal, S_(RC). In step 708, the SWBBG 410 may recordthe channel weight phase applied to the mixer 408 that correspond to thehighest baseband combined channel estimates determined by the CPP 218during the rotation operation. In step 710, the SWBBG 410 may apply therecorded phase as the channel weight to modify the portion of thetransmitted signal received by the additional receive antennas. In endstep 712, the SWBBG 410 may wait to determine new channel weights whenthe operation is periodic or the SWBBG 410 may begin again in step 702to determine new channel weights when the operation is continuous. Theexemplary steps described in FIG. 7 may be extended to the SC MIMO casewith or without support for CL transmit antenna weights.

Another embodiment of the invention may provide a machine-readablestorage, having stored thereon, a computer program having at least onecode section executable by a machine, thereby causing the machine toperform the steps as described above for weight determination in an SCMIMO system for WCDMA/HSDPA.

The approach described herein for weight determination in a SC MIMOsystem for WCDMA/HSDPA may provide a good compromise betweenimplementation complexity and performance gains in the design andoperation of MIMO systems.

Accordingly, the present invention may be realized in hardware,software, or a combination of hardware and software. The presentinvention may be realized in a centralized fashion in at least onecomputer system, or in a distributed fashion where different elementsare spread across several interconnected computer systems. Any kind ofcomputer system or other apparatus adapted for carrying out the methodsdescribed herein is suited. A typical combination of hardware andsoftware may be a general-purpose computer system with a computerprogram that, when being loaded and executed, controls the computersystem such that it carries out the methods described herein.

The present invention may also be embedded in a computer programproduct, which comprises all the features enabling the implementation ofthe methods described herein, and which when loaded in a computer systemis able to carry out these methods. Computer program in the presentcontext means any expression, in any language, code or notation, of aset of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform.

While the present invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiment disclosed, but that the present invention willinclude all embodiments falling within the scope of the appended claims.

1. A method for handling wireless communication, the method comprising: determining a first model of single channel (SC) communication signals received in a first receive antenna and at least one additional model of SC communication signals received in at least one additional receive antenna; combining said first model and said at least one additional model to determine a combined signal component and a combined noise component; determining a plurality of signal strength parameters by maximizing a function based on said combined signal component and said combined noise component for a plurality of phase and amplitude factors, wherein the maximizing the function is performed using a step-based search algorithm; and adjusting a phase and an amplitude factor of at least a portion of said received SC communication signals at each of said at least one additional receive antennas based on a highest value of said determined plurality of signal strength parameters.
 2. The method according to claim 1, comprising determining said first model based on at least one channel estimate and at least one noise estimate that correspond to said first receive antenna.
 3. The method according to claim 2, comprising determining said first model based on at least one interference channel estimate that corresponds to an interference signal.
 4. The method according to claim 1, comprising determining each of said at least one additional model based on at least one channel estimate and at least one noise estimate that correspond to said at least one additional receive antenna.
 5. The method according to claim 4, comprising determining said at least one additional model based on at least one interference channel estimate that corresponds to an interference signal.
 6. The method according to claim 1, wherein each of said at least one additional model comprises an amplitude factor and a phase factor.
 7. The method according to claim 1, comprising determining a plurality of signal-to-noise ratios (SNR) based on said combined signal component and said combined noise component.
 8. The method according to claim 1, comprising determining a plurality of signal-to-interference-and-noise ratios (SINR) based on said combined signal component and said combined noise component.
 9. The method according to claim 1, herein using the step-based search algorithm includes determining said plurality of signal strength parameters over a plurality of phase factor steps in a 360-degrees phase rotation.
 10. The method according to claim 1, wherein using the step-based search algorithm includes determining said plurality of signal strength parameters over a plurality of amplitude factor values in an amplitude range.
 11. A system for handling wireless communication, the system comprising: a single weight baseband generator (SWBBG) that enables determination of a first model of single channel (SC) communication signals received in a first receive antenna and at least one additional model of SC communication signals received in at least one additional receive antenna; said SWBBG enables combining of said first model and said at least one additional model to determine a combined signal component and a combined noise component; said SWBBG enables determination of a plurality of signal strength parameters by maximizing a function based on said combined signal component and said combined noise component for a plurality of phase and amplitude factors, wherein the maximizing the function is performed using a step-based search algorithm; and said SWBBG enables adjustment of a phase and an amplitude factor of at least a portion of said received SC communication signals at each of said at least one additional receive antennas based on a highest value of said determined plurality of signal strength parameters.
 12. The system according to claim 11, wherein said SWBBG determination of said first model based on at least one channel estimate and at least one noise estimate that correspond to said first receive antenna.
 13. The system according to claim 12, wherein said SWBBG enables determination of said first model based on at least one interference channel estimate that corresponds to an interference signal.
 14. The system according to claim 11, wherein said SWBBG enables determination of each of said at least one additional model based on at least one channel estimate and at least one noise estimate that correspond to said at least one additional receive antenna.
 15. The system according to claim 14, wherein said SWBBG enables determination of said at least one additional model based on at least one interference channel estimate that corresponds to an interference signal.
 16. The system according to claim 11, wherein each of said at least one additional model comprises an amplitude factor and a phase factor.
 17. The system according to claim 11, wherein said SWBBG enables determination of a plurality of signal-to-noise ratios (SNR) based on said combined signal component and said combined noise component.
 18. The system according to claim 11, wherein said SWBBG enables determination of a plurality of signal-to-interference-and-noise ratios (SINR) based on said combined signal component and said combined noise component.
 19. The system according to claim 11, wherein the step-based search algorithm enables determination of said plurality of signal strength parameters over a plurality of phase factor steps in a 360-degrees phase rotation.
 20. The system according to claim 9, wherein the step-based search algorithm said SWBBG enables determination of said plurality of signal strength parameters over a plurality of amplitude factor values in an amplitude range.
 21. The system according to claim 11, wherein said SWBBG enables determination of said first model and said at least one additional model after receiving a signal from a channel estimator.
 22. The system according to claim 11, wherein said SWBBG enables generation of a signal that indicates that said channel weight for each of said at least one additional receive antennas has been determined.
 23. The system according to claim 11, wherein said SWBBG comprises a channel estimator, a channel weight generator, a phase and amplitude controller, a phase rotator start controller, and a delay signal generator. 